This application is related to copending and commonly assigned application Ser. No. 417286, entitled "Digitally Controlled Transversal Equalizer", filed Sept. 13, 1982.
This invention is directed to a transversal equalizer and, more particularly, to adaptive equalizers. The invention is specifically directed to an implementation of an adaptive transversal equalizer wherein the transversal equalizer is implemented at IF and its adaptive algorithm is controlled by a digitally based controller in accordance with information derived from the baseband signal.
Conventional adaptive equalizers have utilized a baseband implementation of the transversal equalizer, e.g. a tapped delay line, in addition to baseband processing of the adaptive algorithm. FIG. 1 is a schematic diagram of a 5-tap implementation of a conventional adaptive transversal equalizer, with FIG. 2 illustrating the same equalizer in block diagram form. As shown in FIG. 1, the conventional adaptive transversal equalizer for use in high speed PSK or QPSK applications, receives a demodulated I channel signal at terminal 10 and a demodulated Q channel signal at terminal 20. Each data channel is then delayed in symbol time increments, e.g. the I channel is delayed in a tapped delay line comprising a plurality of delays 12, 14, 16 and 18, and the various delayed signals are then weighted in multipliers 11, 13, 15, 17 and 19 with respective coefficients Al-A5. The weighted signals are then combined in a summer 30. In addition to inputs weighted in accordance with the direct coupled coefficients A-A5, the summer 40 rceives additional inputs from multipliers 50, 52, 54, 56 and 58 which provide weighted samples from the Q channel, these samples being weighted in accordance with cross-coupled coefficients C1-C5. The summer 40 provides an output error signal e to the predictor 60 where the error signals are quantized in a level discriminator 62 which is shown more clearly in FIG. 1A, the series of quantized output signals from the level discriminator 62 are integrated in a low pass filter (LPF) 64, and the integrated error signal is then provided as one input to summing amplifier 66 which adds the integrated error signals to the present error signal to provide a predicted value output at node 68. The predicted value is supplied, with appropriate gain control, as one input to each of mixers 70, 72, 74, 76 and 78 where it is mixed with a respective delayed I channel signal. The mixer output signals are sampled in samplers 80, 82, 84, 86 and 88 and filtered in filters 90, 92, 94, 96 and 98, with the filter outputs providing the appropriate direct coupled coefficients A1-A5 to the multipliers 30-38. The predicted signal value at node 68 is also provided with appropriate gain control to a plurality of mixers for combination with various delayed values of the demodulated Q channel signal, with subsequent sampling, filtering and feeding back of the cross coupled coefficients C1-C5 to the appropriate multipliers 50-58 in a similar manner.
A second summer 41 receives weighted inputs from both the I and Q channels in a similar and provides an error signal output to a second predictor 61 which operates in the same manner as the predictor 60. For a Quadrature Phase Shift Keying (QPSK) modem. The weighted outputs of one channel, including the center weight, are summed with the corresponding weighted outputs of the other channel, excluding the center weight. The output of each summation network is applied to a predictor and comparator, and the two outputs of each predictor and comparator which constitute the errors are applied to the control algorithm of both channels. The control algorithm cross-correlates the output of the transversal equalizer and produces the control signal to change the weights of the transversal equalizer taps.
A problem with such an arrangement is that the functional complexity of the conventional baseband adaptive equalizer illustrated in FIG. 1 renders it difficult to implement and very costly. For example, the multipliers, e.g. multipliers 30-38 and 50-58, used to form the final signal must operate over a frequency range from DC to the maximum frequency of the input signal with good phase characteristics. These multipliers must also operate in all four quadrants, and be reasonably linear for good reduction of intersymbol interference. The error correlators, e.g. 70-78, must satisfy the same requirements, except that linearity requirements are relaxed, since the non-linearity will only affect the convergence time. For high speed QPSK modems operating in the TDMA mode, the conventional baseband equalizer cannot respond fast enough to the variations of successive bursts. These limitations are due to the functional and operational requirements as well as the frequency limitations of components. For example, analog summation amplifiers and four-quadrant amplifiers for high speed operations (&gt;120 Mbit) are not readily available. Although the analog summation or differential amplifiers may be custom built, the four-quadrant multipliers are even difficult to custom build for operations at 120 Mbit/s. Also, for a 120 Mbit QPSK modem, summation networks are not readily available, especially those requiring more than three input summations. Phase linearity on all amplifiers is also important.
A further problem with the conventional equalizer design is that integration of the error signal in the predictor 60 is carried out continuously and the coefficients converge to obtain a minimum mean square error averaged over the entire signal. This technique is very costly at high speeds. Further, since continuous integration of the error function is used, the design is unsuitable for discontinuous burst communications encountered in TDMA applications.